Multi-band transmit/receive feed utilizing PCBS in an air dielectric diplexing assembly

ABSTRACT

In one example an apparatus is provided. The apparatus includes a low frequency radiator, a high frequency radiator, a high frequency waveguide that carries high frequency bands to the high frequency radiator, a low frequency coaxial waveguide coupled to the high frequency waveguide in a coaxial structure, wherein the low frequency coaxial waveguide carries low frequency bands to the low frequency radiator and a low frequency combiner in communication with the low frequency coaxial waveguide, wherein the low frequency combiner comprises a circular low frequency waveguide and air dielectric transmission lines formed by air channels formed above and below a plurality of printed circuits in a metal housing.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims priority to U.S. Provisional Patent ApplicationSer. No. 62/333,519, filed May 9, 2016, which is herein incorporated byreference in its entirety.

The present disclosure relates generally to antennas and, moreparticularly, to a multi-band transmit and receive feed utilizingprinted circuit boards in an air dielectric duplexing assembly.

BACKGROUND

With increasing demand for communication capacity, “multiple satellites”and “multi-band satellites” have been deployed at or very near the sameorbital location (within 0.5 deg of one another). This is commonlyreferred to as “co-locating” the satellite services at a singlesatellite location. This has occurred at numerous satellite locationsaround the globe. Many of these “co-located” satellites are designed totransmit and/or receive large amounts of data (bandwidth) to and/or fromground systems.

The “co-located” satellites are used for both video and internetservices for businesses, homes, and other applications. Co-locatingsatellite capacity has increased in popularity due to limitations ofavailable satellite orbital locations and a desire to deploy singleantenna ground stations that can serve multiple functions (includingbroad band video and internet services).

SUMMARY

According to aspects illustrated herein, there is provided an apparatus,comprising a low frequency radiator, a high frequency radiator, a highfrequency waveguide that carries high frequency bands to the highfrequency radiator, a low frequency coaxial waveguide coupled to thehigh frequency waveguide in a coaxial structure, wherein the lowfrequency coaxial waveguide carries low frequency bands to the lowfrequency radiator and a low frequency combiner in communication withthe low frequency coaxial waveguide, wherein the low frequency combinercomprises a circular low frequency waveguide and air dielectrictransmission lines formed by air channels formed above and below aplurality of printed circuits in a metal housing.

BRIEF DESCRIPTION OF THE DRAWINGS

The teachings of the present invention can be readily understood byconsidering the following detailed description in conjunction with theaccompanying drawings, in which:

FIG. 1 illustrates a side view of an example multi-band transmit/receivefeed of the present disclosure;

FIG. 2 illustrates a top view of the example multi-band transmit/receivefeed of the present disclosure;

FIG. 3 illustrates a bottom view of the example multi-bandtransmit/receive feed of the present disclosure;

FIG. 4 illustrates an exploded side view of the example multi-bandtransmit/receive feed of the present disclosure;

FIG. 5 illustrates an exploded isometric view of the example multi-bandtransmit/receive feed of the present disclosure;

FIG. 6 illustrates a top view of an example printed circuit board (PCB)of the present disclosure;

FIG. 7 illustrates a top view of a plurality of PCBs inside of theexample multi-band transmit/receive feed of the present disclosure;

FIG. 8 illustrates a side view of the example multi-bandtransmit/receive feed of the present disclosure without a metal housingaround a low frequency combiner;

FIG. 9 illustrates an isometric view of the example multi-bandtransmit/receive feed of the present disclosure with channel wallsextended in the “z” direction;

FIG. 10 illustrates a side view of the example multi-bandtransmit/receive feed of the present disclosure enclosed with a metalhousing an coupled to a dual polarity transceiver;

FIG. 11 illustrates a side view of the example multi-bandtransmit/receive feed of the present disclosure in a reflector antennasystem;

FIG. 12 illustrates a first example air dielectric transmission line;

FIG. 13 illustrates a second example air dielectric transmission line;

FIG. 14 illustrates a third example air dielectric transmission line;

FIG. 15 illustrates a fourth example air dielectric transmission line;

FIG. 16 illustrates a fifth example air dielectric transmission line;

FIG. 17 illustrates a sixth example air dielectric transmission line;

FIG. 18 illustrates a seventh example air dielectric transmission line;

FIG. 19 illustrates an eighth example air dielectric transmission line;

FIG. 20 illustrates a ninth example air dielectric transmission line;

FIG. 21 illustrates an isometric view of an assembly of a tenth exampleair dielectric transmission line;

FIG. 22 illustrates an eleventh example air dielectric transmissionline;

FIG. 23 illustrates a twelfth example air dielectric transmission line;

FIG. 24 illustrates a top view of the example multi-bandtransmit/receive feed of the present disclosure with two KU waveguideports;

FIG. 25 illustrates an example circular polarizer mechanism thatsupports two low band circular polarities;

FIG. 26 illustrates an isometric view of the example circular polarizermechanism coupled to the low frequency combiner;

FIG. 27 illustrates a second example circular polarizer mechanism thatsupports two low band circular polarities;

FIG. 28 illustrates a top view of the plurality of PCBs;

FIG. 29 illustrates a side view of the plurality of PCBs and a z heightseparation between the plurality of PCBs;

FIG. 30 illustrates a graph of phase differential between orthogonalhorizontal and vertical linear polarity components;

FIG. 31 illustrates views of an example micro-strip of the presentdisclosure;

FIG. 32 illustrates a second graph of phase differential betweenorthogonal horizontal and vertical linear polarity components; and

FIG. 33 illustrates a third graph of phase differential betweenorthogonal horizontal and vertical linear polarity components.

To facilitate understanding, identical reference numerals have beenused, where possible, to designate identical elements that are common tothe figures.

DETAILED DESCRIPTION

The present disclosure relates to a multi-band transmit/receive feedutilizing printed circuits in an air dielectric diplexing assembly. Asdiscussed above, co-locating satellite capacity has increased inpopularity due to limitations of available satellite orbital locationsand a desire to deploy single antenna ground stations that can servemultiple functions (including broad band video and Internet services).

One solution for ground systems is to use multiple ground antennasystems at a given ground location each designed for one, or maybe two,frequency bands. However, that is not desirable from a marketing(aesthetic), zoning, or cost perspective. So there is considerabledemand for a single ground antenna system (as described in thisdisclosure) that is capable of receiving and transmitting all of thewanted bands from and to collocated satellites.

This gives rise to significant design challenges for the ground antennasystem. The ground antenna system must transmit and receive broad bandwidths of data contained in multiple frequency bands to one satellitelocation. In addition, most of these signals are carried in twopolarities. The ground systems must be able to combine (together) and/ordiplex (separate) the various frequency bands and polarities. As withany satellite earth station system this must be accomplished in such amatter to minimize unwanted interference from and to other neighboringsatellites.

In search of a solution several approaches have been developed in recentyears but have had limited or no commercial success due to their cost,difficulty to manufacture, and/or compromised performance. The presentdisclosure addresses these issues.

FIG. 1 illustrates a side view of an example multi-band transmit/receivefeed 100 of the present disclosure. In one embodiment, the multi-bandtransmit/receive feed 100 includes a low frequency radiator 102, a highfrequency radiator 104, a high frequency waveguide 106, a coaxialstructure 108, a low frequency combiner 110 and a circular low frequencywaveguide 128. Together these elements form a diplexer that separateslow frequency band signals from high frequency band signals.

In one embodiment, the high frequency band signals (e.g., high frequencyband A having a range of approximately 29.5 gigahertz (GHz) to 30 GHzand high frequency band B having a range of approximately 19.7 GHz to20.2 GHz) are diplexed to and/or from a high frequency interface port114. The low frequency band signals (e.g., low frequency band C having arange of approximately 10.7 GHz to 12.75 GHz) are transmitted to and/orfrom a low frequency interface port 112.

In one embodiment, the coaxial structure 108 may comprise the highfrequency waveguide 106 and a low frequency coaxial waveguide 116. Thehigh frequency waveguide 106 may be a small inner circular waveguide forhigh frequency bands. The low frequency coaxial waveguide 116 may belocated around the high frequency waveguide 106 (e.g., in an annulararrangement) to form the coaxial structure 108. The low frequencycoaxial waveguide 116 may be used for low frequency bands.

The relatively long small diameter of the high frequency waveguide 106may carry high frequency bands between the high frequency interface port114 and the high frequency radiator 104. In one embodiment, the highfrequency radiator 104 may be a dielectric radiator, dielectric horn, orsimple waveguide or horn. The high frequency waveguide 106 can be loadedwith a dielectric material in order to reduce the waveguide size (e.g.,a diameter) and/or to improve and enhance radiating characteristics.

The low frequency coaxial waveguide 106 may carry both polarities oflower frequency bands between the low frequency radiator 102 and the lowfrequency combiner 110. In one embodiment, the low frequency radiator102 may be a circular corrugated horn.

In one embodiment, the low frequency combiner 110 may be enclosed by ametal housing that comprises a top metal housing portion 120 and abottom metal housing portion 122. FIG. 8 illustrates a side view of thelow frequency combiner 110 without the metal housing. FIG. 2 illustratesa top view of the example multi-band transmit/receive feed 100 of thepresent disclosure and FIG. 3 illustrates a bottom view of the examplemulti-band transmit/receive feed 100 of the present disclosure.

FIG. 4 illustrates an exploded side view of the multi-bandtransmit/receive feed 100. In one embodiment, the low frequency combiner110 may include a first printed circuit board (PCB) 130 and a second PCB132. Although FIG. 4 illustrates the printed circuits on a PCB, itshould be noted that the circuits may be printed on any type of mediumincluding thin films commonly used in flexible printed circuits. Thus,the term PCB may generically refer to printed circuits that are printedon any type of medium including thin films.

In one embodiment, the metal housing may also include a middle metalhousing portion 124. The first PCB 130 may be enclosed by the top metalhousing portion 120 coupled to the middle metal housing portion 124. Thesecond PCB 132 may be enclosed by the middle metal housing portion 124coupled to the bottom metal housing portion 122.

FIG. 5 illustrates an exploded isometric view of the multi-bandtransmit/receive feed 100. The exploded isometric view in FIG. 5illustrates a first trace 162 and a second trace 164 in the first PCB130 and a third trace 166 and a fourth trace 168 in the second PCB 132.The first trace 162 is contained within a volume of a first channel 142(illustrated in FIG. 7) formed by the metal housings 120 and 124. Thesecond trace 164 is contained within a volume of a second channel 144(illustrated in FIG. 7) formed by the metal housings 120 and 124. Thethird trace 166 is contained within a volume of a third channel 146(illustrated in FIG. 7) formed by the metal housings 124 and 122. Thefourth trace 168 is contained within a volume of a fourth channel 148(illustrated in FIG. 7) formed by the metal housings 124 and 122. Itshould be noted that the terms “first,” “second,” “third,” and “fourth”are used purely as labels to differentiate between the differentchannels. The terms “first,” “second,” “third,” and “fourth” are notintended to connote any order or sequence.

In one embodiment, the low frequency combiner 110 may carry lowfrequency bands between the low frequency coaxial waveguide 116 and thecircular low frequency waveguide 128. The circular low frequencywaveguide 128 may carry the low frequency bands to and/or from the lowfrequency interface port 112. In one embodiment, the first PCB 130 maybe positioned between the top metal housing portion 120, the middlemetal housing portion 124, while the second PCB 132 may be positionedbetween the middle metal housing portion 124 and the bottom metalhousing portion 122, as noted above.

FIG. 6 illustrates an example of the first PCB 130. It should be notedthat the second PCB 130 may have similar features as the first PCB 130.In one embodiment, the first PCB 130 may include the first metal trace162 and the second metal trace 164. In one embodiment, the first PCB 130may also include a plurality of platted through vias 150 that align withthe walls of an area that is associated with the first channel 142 andthe second channel 144. In other words, the plurality of platted throughvias 150 may follow a perimeter or an outline of the first channel 142and the second channel 144.

In one embodiment, the PCB board of the first PCB 130 may extend outsideof the first channel 142 and the second channel 144 as shown in FIG. 6.The platted through vias 150 may align with the walls of the firstchannel 142 and the second channel 144 to block and contain the lowfrequency band effectively. As a result, the plurality of plattedthrough vias 150 may electrically extend the side walls of the firstchannel 142 and the second channel 144 in a “z” direction (e.g., out ofthe page in FIG. 6) to further constrain the signal path to within theboundaries of the side wall.

In some embodiments, the plurality of platted through vias 150 may beused to connect a duplicate trace of the first metal trace 162 and thesecond metal trace 164 that are located on an opposite side of the firstPCB 130. Similarly, the second PCB 132 may also include the plurality ofplatted through vias 150 that may be used to connect a duplicate traceof the third metal trace 166 and the fourth metal trace 168 that arelocated on the opposite side of the second PCB 132. An example of thetraces on both sides of a printed circuit board that are connected byvias is shown in FIG. 12, and discussed in further detail below.

In some embodiments, the portions of the first PCB 130 around the areaassociated with the first channel 142 and the second channel 144 andportions of the second PCB 132 around the area associated with the thirdchannel 146 and the fourth channel 148 can be eliminated while extendingthe channel walls in the “z” direction. As a result, the plurality ofplatted through vias 150 may be eliminated. Extending the channel wallsin the “z” direction may be convenient for visualization of the signalpath constraints and provide more efficient High Frequency StructureSimulator (HFSS) modeling of radio frequency (RF) performance.

FIG. 9 illustrates an isometric view of an example multi-bandtransmit/receive feed 900 having a channel wall 902 and 904 extended toclose the channels, as described above. FIG. 9 also illustrates thefirst channel 142, the second channel 144, the third channel 146 and thefourth channel 148 formed by the housings that enclose the metal traces.

FIG. 7 illustrates a top view of the first PCB 130 and the second PCB132 inside of the example multi-band transmit/receive feed 100. FIG. 7illustrates the second PCB 132 having a third trace 166 in the thirdchannel 146 and a fourth trace 168 in a fourth channel 148 of the secondPCB 132. The second PCB 132 may also have a plurality of platted throughvias 150 as shown in FIG. 7. FIG. 7 also illustrates the first channel142, the second channel 144, the third channel 146, and the fourthchannel 148 formed by the housings around the first metal trace 162, thesecond metal trace 164, the third metal trace 166, and the fourth metaltrace 168, respectively.

Referring back to FIG. 5, in one embodiment, the first PCB 130 enclosedbetween the top metal housing portion 120 and the middle metal housingportion 124 may have hollowed out air channels above and below the firstmetal trace 162 and the second metal trace 164, such that the first PCB130, the top metal housing portion 120 and the middle metal housingportion 124 together form air dielectric transmission lines. Similarly,the second PCB 132 enclosed between the middle metal housing portion 120and the bottom metal housing portion 122 may have hollowed out airchannels above and below the third metal trace 166 and the fourth metaltrace 168, such that the second PCB 132, the middle metal housingportion 124 and the bottom metal housing portion 122 together may alsoform air dielectric transmission lines. For example, the air dielectrictransmission lines may be formed by metal conductors (possibly printedcircuit board traces) contained within air filled metal channelsresulting in a low loss transmission line. Various differentconfigurations of the air dielectric transmission lines are discussed infurther detail below with reference to FIGS. 12-26.

In one embodiment, the first PCB 130 may be for a first polarity of thelow frequency bands. In one embodiment, the second PCB 132 may be for asecond polarity of the low frequency bands.

Referring back to FIG. 7, in one embodiment, the first PCB 130 for thefirst polarity has two signal paths supported by the first metal trace162 and the second metal trace 164 contained in the first channel 142and the second channel 144, respectively. On one end, the first metaltrace 162 and the second metal trace 164 may enter the low frequencycoaxial waveguide 116 on opposing sides using a first probe 152 and asecond probe 154. On the other end the first metal trace 162 and thesecond metal trace 164 may enter the circular low frequency waveguide128 on opposing sides with a third probe 172 and a fourth probe 174. Inone embodiment, the first metal trace 162 and the second metal trace 164may be electrically equal in length (e.g., an equal phase length) inorder to preserve symmetric field excitation inside both the lowfrequency coaxial waveguide 116 and the circular low frequency waveguide128.

Similarly, the second PCB 132 for the second polarity (which isorthogonal to the first polarity) has two signal paths supported by thethird metal trace 166 and the fourth metal trace 168 contained in thethird channel 146 and the fourth channel 148, respectively. On one end,the third metal trace 166 and the fourth metal trace 168 may enter thelow frequency coaxial waveguide 116 on opposing sides using a fifthprobe 156 and a sixth probe 158. On the other end, the third metal trace166 and the fourth metal trace 168 may enter the circular low frequencywaveguide 128 on opposing sides with a seventh probe 176 and an eighthprobe 178. In one embodiment, the third metal trace 166 and the fourthmetal trace 168 may be electrically equal in length (e.g., equal phaselength) in order to preserve symmetric field excitation inside both thelow frequency coaxial waveguide 116 and the circular low frequencywaveguide 128.

In one embodiment, the first probe 152 and the second probe 154 may belocated approximately 90 degrees relative to the fifth probe 156 and thesixth probe 158. Said another way, a line drawn between the first probe152 and the second probe 154 may be perpendicular to a line drawnbetween the fifth probe 156 and the sixth probe 158. In one embodiment,the first probe 152, the second probe 154, the fifth probe 156 and thesixth probe 158 may be arranged around a perimeter of an opening. In oneembodiment, the coaxial structure 108 may be located through the openingformed by the first probe 152, the second probe 154, the fifth probe 156and the sixth probe 158.

In one embodiment, the third probe 172 and the fourth probe 174 may belocated approximately 90 degrees relative to the seventh probe 176 andthe eighth probe 178. Said another way, a line drawn between the thirdprobe 172 and the fourth probe 174 may be perpendicular to a line drawnbetween the seventh probe 176 and the eighth probe 178. In oneembodiment, the third probe 172, the fourth probe 174, the seventh probe176 and the eighth probe 178 may be arranged around a perimeter of anopening. In one embodiment, the circular low frequency waveguide 128 maybe located through the opening formed by the third probe 172, the fourthprobe 174, the seventh probe 176 and the eighth probe 178.

In one embodiment, the first probe 152, the second probe 154, the thirdprobe 172, the fourth probe 174, the fifth probe 156, the sixth probe158, the seventh probe 176 and the eighth probe 178 may be printedprobes. In other words, the first probe 152, the second probe 154, thethird probe 172, the fourth probe 174, the fifth probe 156, the sixthprobe 158, the seventh probe 176 and the eighth probe 178 may be printedinto the first PCB 130 and the second PCB 132, respectively.

Although the cross-sections of the high frequency waveguide 106, thehigh frequency interface port 114, the low frequency waveguide 128 andthe low frequency interface port 112 are shown in FIGS. 1-8 as beingcircular, it should be noted that the cross-sections may have a widevariety of shapes. Other example shapes of the cross sections of thehigh frequency waveguide 106, the high frequency interface port 114, thelow frequency waveguide 128 and the low frequency interface port 112 mayinclude an ellipse, a rectangular, a square, a hexagon, an octagon, anynumber of polygon, or a wide variety of oblong shapes. Furthermore theshape and/or aspect ratio of the cross section can change along thelength of the waveguide.

In addition, although the cross-section of the outer diameter of the lowfrequency coaxial waveguide 116 is shown in FIGS. 1-8 as being circular,it should be noted that the cross section can be a wide variety ofshapes. Example shapes of the cross section of the low frequency coaxialwaveguide 116 may include an ellipse, a rectangular, a square, ahexagon, an octagon, any number of polygon, or a wide variety of oblongshapes. Furthermore the shape and/or aspect ratio of the cross sectioncan change along the length of the waveguide.

FIG. 10 illustrates a side view of the multi-band transmit/receive feed100 of the present disclosure enclosed with a metal housing 1008 and alens 1006. The multi-band transmit/receive feed 100 may be coupled to adual polarity transceiver 1002.

In one embodiment, the circular low frequency waveguide 128 of the lowfrequency combiner 110 may support both polarities of the low frequencyband C that can be connected to a variety of device. One example devicemay be a dual polarity low noise block (LNB) down converter 1004 that isshown coupled to the low frequency combiner 110 in FIG. 10. The dualpolarity LNB down converter 1004 may convert low frequency band Csignals to an even lower frequency band F.

FIG. 11 illustrates a side view of the multi-band transmit/receive feed100 in a reflector antenna system 1100. In one embodiment, the reflectorantenna system 1100 may include the multi-band transmit/receive feed100, a reflector 1104 and a satellite receiver/modern set top box 1102.As noted above, the dual polarity LNB down converter 1004 may convert alow frequency band C signal 1110 (e.g., 10.7 GHz-12.75 GHz) into a lowerfrequency F band signal. The converted lower frequency F band signal maybe sent through low cost long coaxial cables 1108 from the set top box1102 to the set top box 1102.

In one embodiment, the high frequency waveguide 106 may support bothpolarities of a high frequency band B signal 1112 (e.g., 19.7 GHz-20.2GHz) and a high frequency band A signal 1114 (e.g., 29.5 GHz-30 GHz).The high frequency waveguide 106 may be connected to a variety ofdevices including the dual polarity transceiver 1002. The dual polaritytransceiver 1102 may convert the high frequency band B signal 1112 to avery low frequency band D signal that is sent through a low cost longcoaxial cable 1106 to the set top box 1102.

The dual polarity transceiver 1102 may also convert a lower frequencyband E signal coming from the set top box 1102 into the high frequencyband A signal 1114. The high frequency band A signal 1114 may be sentfrom the dual polarity transceiver to the high frequency waveguide 114.

Several embodiments of the present disclosure provide excellent broadband linear polarity performance. For example, if the probes in the LNB1004 are aligned with the probes 172, 174, 176, 178 in the low frequencywaveguide 128, then the LNB 1004 efficiently receives both horizontaland vertical linear polarity signals.

Several embodiments of the present disclosure provide excellent broadband circular polarity performance. If the probes in the LNB 1004 areoriented at 45 degrees relative to the probes 172, 174, 176, 178 in thelow frequency waveguide 128, then the LNB 1004 can efficiently receiveboth Right Hand and Left Hand circular polarity signals, provided thatthe distance z between the traces (and probes) on the two boards and thelengths of the traces 162, 164, 166, 168 are appropriately sized inorder to create the needed 90 degree phase differential betweenorthogonal linear components that define each circular polarity signal.In other words, a length of the channels within each one of theplurality of printed circuits and/or a distance along a z-axis betweentwo or more of the plurality of circuits may be set in combination withan additional circular polarizer mechanism comprising one of a varietyof different shapes, as discussed below, to receive and/or transmitcircular polarity signals.

Furthermore, understanding the phase differential vs frequency responsefor each of the components and properly combining appropriately sizedcomponents can be used to provide extremely good circular polarityperformance (e.g., low cross polarity levels) by maintaining 90 degreephase differential over significant frequency band width. In particular,the sizes can be chosen such that the phase differential (versusfrequency response) between orthogonal linear components introduced inthe coaxial and low frequency waveguides by the z distance d₅ (e.g., asshown in FIG. 29 and discussed below) between the boards is oppositelysloped compared to the phase differential (vs. frequency response)introduced by the difference in trace lengths.

FIGS. 12-26 illustrate different example designs of the air dielectricchannels in the low frequency combiner 110 formed by the PCBs 130 and132 and the metal housing portions 120, 122 and 124. The referencenumerals have been re-numbered in FIGS. 12-26 for ease of explanation.However, the channel walls may correspond to the metal housing portions120, 122 and 124 and the boards may refer to either the PCB 130 or 132.

FIG. 12 illustrates an isometric view 1202 and a cross-sectional view1203 of a first example air dielectric channel. The first example airdielectric channel may be an air dielectric stripline having a doublesided trace with vias. In one embodiment, the air dielectric channel maybe formed by a board dielectric 1212 enclosed by a top channel wall1204, a bottom channel wall 1206, a left side wall 1208 and a right sidewall 1210.

In one embodiment, the top channel wall 1204 and the bottom channel 1206may be closer to the top trace 1216 and the bottom trace 1218,respectively, then the left side wall 1208 and the right side wall. Thefields may be contained in an air dielectric 1220 formed by the volumebetween the sides of the board dielectric 1212 and the top channel wall1204, the bottom channel wall 1206, the left side wall 1208 and theright side wall 1210.

In one embodiment, the board dielectric 1202 may include one or morevias 1214. The board dielectric 1212 may include a top trace 1216 and abottom trace 1218 on opposing sides of the board dielectric 1212. Thevias 1214 may be formed through the board dielectric 1212 and used toconnect the top trace 1216 and the bottom trace 1218 to effectivelycreate a single thick conductor/trace. In one embodiment, a thickness ofthe board dielectric 1212 may be 0.20 inches (in) or greater or athickness of 0.003 in or less. As a result, even when using conventionallow cost board material, losses may be reduced by eliminating fields inthe air dielectric 1220 directly between the top trace 1216 and thebottom trace 1218 between the line of vias 1214 on either side of thetop trace 1216 and the bottom trace 1218. As a result, most of thefields may be concentrated in the low loss air dielectric 1220.

FIG. 13 illustrates an isometric view 1302 and a cross-sectional view1303 of a second example air dielectric channel. The second example airdielectric channel may be an air dielectric stripline with a singlesided trace. The second example air dielectric channel may besubstantially similar to the first example air dielectric channel exceptthat a single trace 1318 may be used instead of two on each side of thedielectric board 1312.

FIG. 13 illustrates how a single sided trace can be also used resultingin the field between the single trace 1318 and a bottom channel wall1306 of the channel passing through all air of the low loss airdielectric 1320. In contrast, the fields between the single trace 1318and the top channel wall 1304 pass through both the low loss airdielectric 1320 and the dielectric board 1312, which may slightlyincrease loss. However, this loss can be reduced if a thickness of thedielectric board 1312 is reduced so the fields pass through lessdielectric as shown in FIG. 14.

FIG. 14 illustrates an isometric view 1402 and a cross-sectional view1403 of a third example air dielectric channel. The third example airdielectric channel may be an air dielectric stripline with a singlesided trace and a thin dielectric board. The third example airdielectric may be similar to the second example air dielectric channelexcept that the third example air dielectric may have a thin dielectricboard 1412 with a single trace 1418. For example, the thickness of thethin dielectric board 1412 may be less than 0.0003 in, as noted above.Thus, the design of the third example air dielectric channel may be wellsuited for thin film substrates commonly used in flex circuits.

FIGS. 12-14 illustrate essentially strip-line structures with airdielectrics where the distances between the traces and the side walls ofthe channel are significantly greater than the distances from the tracesto the top channel wall or the bottom channel wall. FIG. 15 illustratesan isometric view 1502 and a cross-sectional view 1503 of a fourthexample air dielectric channel. The fourth example air dielectricchannel may be a primarily air dielectric quasi-stripline with a doublesided trace.

In one embodiment, the air dielectric channel may be formed by a boarddielectric 1512 enclosed by a top channel wall 1504, a bottom channelwall 1506, a left side wall 1508 and a right side wall 1510. The boarddielectric 1512 may include a first trace 1516 and a second trace 1518coupled to the board dielectric 1512 on opposing sides of the boarddielectric 1512.

In the embodiment illustrated in FIG. 15, a distance d₁ between the leftside wall 1508 or the right side wall 1510 and the first trace 1516 orthe second trace 1518 may be approximately equal to a distance d₂between the top channel wall 1504 or the bottom channel wall 1506 andthe first trace 1516 or the second trace 1518. As a result, the fieldsmay be distributed more uniformly in an air dielectric 1520 all aroundthe first trace 1516 and the second trace 1518 forming a primarily airdielectric quasi-stripline transmission line with a small portion of thefields in the board dielectric 1512 on either side of the first trace1516 or the second trace 1518.

FIG. 16 illustrates an isometric view 1602 and a cross-sectional view1603 of a fifth example air dielectric channel. The fifth example airdielectric channel may be a primarily air dielectric micro-strip with asingle sided trace. FIG. 16 illustrates an asymmetrical placement of thedielectric board 1612 and a single trace 1618. For example, a distancebetween a top channel wall 1604 and the single trace 1618 may be greaterthan a distance between a bottom channel wall 1606 and the single trace1618. FIG. 16 illustrates that even with relatively thick dielectricboards 1612, the loss can be reduced with the asymmetrical placement ofthe dielectric board 1612. For example, the fields may become lessconcentrated in the air dielectric 1620 between the dielectric board1612 and the top channel wall 1604 compared to the concentration of thefields in the air dielectric 1620 between the dielectric board 1612 andthe bottom channel wall 1606.

FIG. 17 illustrates an isometric view 1702 and a cross-sectional view1703 of a sixth example air dielectric channel. The sixth example airdielectric channel may be a primarily air dielectric micro-strip with adouble sided trace. FIG. 17 illustrates an asymmetrically locateddielectric board 1712. The dielectric board 1712 may have a first trace1716 and a second trace 1718 on opposing sides of the dielectric board1712 and one or more vias 1514.

FIG. 18 illustrates an isometric view 1802 and a cross-sectional view1803 of a seventh example air dielectric channel. The seventh exampleair dielectric channel may be an air dielectric with a double sidedtrace. The seventh example air dielectric may have a distance d₃ betweenthe sidewalls and the dielectric board that is less than a distance d₄between the top or bottom channel walls and the dielectric board. Theloss of the seventh example air dielectric channel may increase as thefield concentration in a board dielectric 1812 that is lossy wouldincrease compared to those in FIGS. 12 and 15, for example. Given thebasic structure whether using “single sided traces” or “double sidedtraces with vias” the field concentration (density) between the traceand the channel walls is dependent upon the distance between the traceand the channel walls. The field concentration/density will be greaterbetween the trace and channel wall(s) for those walls that are closer tothe trace.

It should also be noted that other channel cross section shapes (otherthan rectangular) could be used such as elliptical, circular, or anynumber of polygons, to name just a few. FIG. 19 illustrates an isometricview 1902 and a cross-sectional view 1903 of a different cross sectionshape.

In another example, the cross-section of the top channel and the crosssection of the bottom channel that the board and trace sets between canbe differently shaped as show in FIG. 20. FIG. 20 illustrates anisometric view 2002 and a cross-sectional view 2003 of a cross sectionof the top channel and the cross section of a bottom channel havingdifferent shapes. As a result, it should be noted that a wide variety ofchannel shapes can be chosen to implement a low loss primarily airdielectric cost effective transmission line using commonly availablePCB, dielectric boards or flex circuit film or similar shape.

For many of the structures discussed above it should be noted that largeportions of the PCB, or dielectric board, can be removed to furtherreduce loss as shown in FIG. 21. FIG. 21 illustrates various views 2102,2104 and 2106 of the PCB or dielectric board that have large portionsremoved in areas 2108, 2110, 2112 and 2114. In addition, for any of thestructures discussed above it should be noted that one side of thechannel could be closer to the trace than the other if needed for aparticular application as shown in an isometric view 2202 and across-sectional view 2203 in FIG. 22.

FIG. 23 illustrates an alternative to PCB or film. FIG. 23 illustratesan isometric view 2302 and a cross-sectional view 2303 a twelfth exampleair dielectric channel formed by a stamped metal. FIG. 23 illustrates aconductor 2312 formed by a stamped metal that is held in place byplastic supports 2320. FIG. 23 illustrates one example with plasticsupports 2320 between the conductor 2312 and channel walls resulting ina primarily air dielectric stripline structure. Numerous other types ofsupports from any of the channel walls could be used to support thetrace.

The embodiments of the air dielectric channel illustrated in FIGS. 12-23provide a more robust design than conventional rectangular waveguides.The embodiments of FIGS. 12-23 are less susceptible to performancedegradations even due to possible gaps in the side walls where the topand bottom castings mate (e.g., the top metal housing portion 120 andthe middle housing portion 124 or the middle housing portion 124 and thebottom metal housing portion 122). For all of the above embodiments, thefield concentrations leaking into the gaps are relatively small andtherefore do not degrade performance very much. This is because thedirection of the E-field near the gaps in the side walls is primarilyperpendicular to the sidewalls (parallel to the top and bottom wall),making it less likely to leak into the gaps.

Furthermore, for embodiments where the conductor (e.g., the structureformed by the metal traces and the vias that connect the metal traces)is very close to the bottom wall compared to the other walls, the fieldsare concentrated between the conductor and the bottom walls so thatalmost no fields enter the gaps in the side walls. So gaps in the sidewalls degrade performance very little if at all. In fact, the sidewallscould in theory be completely removed and the fields would stayconcentrated between trace and the bottom wall.

For embodiments where the conductor is very close to both the top walland the bottom wall compared to the side walls, the fields may beconcentrated between the conductor and the bottom wall and between theconductor and the top wall so that almost no fields enter the gaps inthe side walls. So gaps in the side walls degrade performance verylittle if at all. In fact, the sidewalls could be completely removed andthe fields would stay concentrated between trace and the bottom wall.

As discussed above, some embodiments use some form of a primarily airfilled low loss transmission line where most of the fields areconcentrated in the air resulting in relatively low loss. However, asshown in FIG. 31, a micro-strip can be used where most of the fields arecontained within the relative higher loss board dielectric, resulting ina relatively high loss transmission line. FIG. 31 illustrates anisometric view 3102 and a cross-sectional view 3103 of a micro-strip.Although the higher loss is generally not desired, the embodimentillustrated in FIG. 31 has two advantages. First, the space, andtherefore the material, needed to support the transmission line can bereduced (e.g., due to the higher dielectric lowering “shortening” thepropagation wavelength. Second, the middle metal housing 124 can beeliminated because the ground plane on the micro-strip boards separatesand isolates the transmission lines and fields for the two polarities.Although not shown, the conventional strip-line could also be usedwhere, for each polarity, all of the fields are contained within theboard dielectric between the centered trace and top and bottom groundplanes.

FIG. 24 illustrates a top view of an example multi-band transmit/receivefeed 2400 with two KU waveguide ports 2402 and 2404. The multi-bandtransmit/receive feed 2400 may have two circular low frequency waveguideports. The KU waveguide port 2402 may be located at the first circularlow frequency waveguide port and the KU waveguide port 2404 may belocated at the second circular low frequency waveguide port.

In one embodiment, the KU waveguide port 2402 may support a first KUpolarity, while the KU waveguide port 2404 may support a second KUpolarity. The advantage of this embodiment is that channels 2406, 2408,2410 and 2412 and respective traces 2416, 2418, 2420 and 2422 between acoaxial waveguide 2430 and the KU waveguide ports 2402 and 2404 can besomewhat shorter than the channels and traces in the embodimentillustrated in FIG. 1. This results in slightly less loss, however itrequires the LNB to have 2 physically separated ports increasing thesize and cost of the LNB.

The embodiments of FIGS. 1 and 24 may support two low band linearpolarities. However, with an addition of a circular polarity polarizermechanism in the KU waveguide between the combiner 110 and LNB 1004 ofthe embodiment of FIG. 1, two low band circular polarities can besupported. If a circular polarity polarizer mechanism is added to bothKU waveguides of the embodiment of FIG. 24 then two low band circularpolarities can be supported. FIG. 25 illustrates an example circularpolarizer mechanism 2500. The circular polarizer mechanism 2500 mayinclude a pair of double ridge polarizers 2502. The pair of double ridgepolarizers may be integrated into either the bottom casting of thecombiner 110 or the LNB 1004 casting as illustrated in FIG. 26.

In one embodiment, the circular polarity polarizer mechanism can bealigned with either the probes 172 and 174, or the probes 176 and 178.In order to receive circular polarity LNB probes are oriented at 45degrees relative to the additional CP mechanism and relative to probes172 and 174, or 176 and 178. This additional mechanism is not typicallyrequired for most applications, but does add another degree of freedomto achieve excellent broad band circular polarity performance.

FIG. 27 illustrates another example of a circular polarizer 2700. Thecircular polarizer 2700 may include a septum structure 2702. Thecircular polarizer 2700 may be coupled to the bottom casting of thecombiner 110 or the LNB 1004 casting similar to the circular polarizer2500 as illustrated in FIG. 26.

It should be noted that FIGS. 25 and 27 illustrate a few examplepolarizer mechanisms or structures that may be implemented. Numerousother polarizer structures could be used including oblong (elliptical,rectangular, or other) waveguides with single, dual, or quad ridges, orsimply oblong waveguides, or oblong dielectric vanes in the axisymmetricor oblong waveguides, or dielectrics that line some, or portions, of thewaveguide walls, to name just a few.

Circular polarity can be defined as the vector sum of two orthogonal (90apart in space) linear components that are also 90 degrees out of phase.Mechanisms that introduce a 90 degree phase lead or lag in oneorthogonal linear component relative to the other orthogonal linearcomponent will convert circular polarity signals to linear polaritysignals that the LNB can receive.

The embodiment in FIGS. 28 and 29 illustrate examples based on varyinglengths of traces 2802, 2804, 2806 and 2808 and a z distance d₅. In oneexample, the lengths of the traces 2802 and 2804 that support linearpolarity (LP) component 1 are equal to the lengths of the traces 2806and 2808 that support the orthogonal LP component 2, so no differentialphase is introduced by the traces 2802, 2804, 2806 and 2808. However,the traces 2802 and 2804 that support the LP component 1 and the traces2806 and 2808 that support the LP component 2 may be separated by thedistance d₅ along the z-axis illustrated in FIG. 29. As a result, the LPcomponent 2 travels the distance d₅ more in the coaxial waveguide whilethe LP component 1 travels the distance d₅ more in the low frequencycircular guide. The phase velocity and corresponding wavelength (λg)inside the circular waveguide changes considerably more as a function ofoperating frequency (f) than it does inside the coaxial waveguide.

In one embodiment, FIG. 30 illustrates a graph 3000 that illustrates aresulting phase differential that may be introduced by the substantiallength of the distance d₅ of the low frequency circular waveguide, thephase differential introduced by the circular polarizer mechanism 2500with the pair of double ridge polarizers 2502, and the resulting sumtotal phase differential. Perfect conversion of circular polarity signalto linear polarity occurs when the sum total phase differential is 90degrees. As shown by the graph 3000, the phase differential versusfrequency that is introduced by the pair of double ridged polarizers2502 may be sloped in the same direction as the phase differentialintroduced by the distance d₅ between the traces 2802 and 2804 thatsupport the LP component 1 and the traces 2806 and 2808 that support theLP component 2. As a result, the resulting sum total phase differentialvs. frequency is very sloped, which severely limits the CP band widthperformance.

As shown in the graph 3000, the sum total phase differential is close to90 degrees over part of the band but near the band edges (e.g., at 10.7GHz and 12.7 GHz) the phase differential is significantly different than90 degrees, which as illustrated in the graph 3000 results in very poorradiated CP Xpol performance over a wide frequency band when combinedwith Horn radiating structures. Most satellite applications require −20to −25 CP Xpol performance. This example provides −20 dB CP Xpol orbetter, for only a relatively narrow frequency band of 11.2 to 11.95GHz, and has very poor CP Xpol of −9.7 dB at 10.7 GHz and −14.7 dB at12.7 GHz.

In various embodiments, changing the length of the traces 2802, 2804,2806 and 2808 and the distance d₅ in FIGS. 28 and 29 may vary theperformance of the multi-band transmit/receive feed 100. For example,the lengths of traces 2802 and 2804 that support linear polaritycomponent 1 may be equal to the lengths of the traces 2806 and 2808 thatsupport linear polarity component 2. As a result, no differential phaseis introduced by the lengths of the traces 2802, 2804, 2806 and 2808. Inaddition, the traces 2802, 2804, 2806 and 2808 may be separated by avery small distance d₅ along the z axis so the resulting phasedifferential introduced by linear polarity component 1 traveling aslightly longer distance than linear polarity component 2 in thecircular wave guide is very small. So in this case the additionalcircular polarity mechanism (ridge polarizer in this example) introducesmost of the needed 90° phase differential. This may lead to a phasedifferential that is close to 90 deg over part of the band, but near theband edges (10.7 GHz, and 12.7 GHz) the phase differential issignificantly different than 90 degrees. This may result in poor CPconversion, specifically substantially degraded CP Xpol performance. Itshould be noted that most conventional CP polarizer mechanisms also havesignificant band width performance limitations.

As noted above, understanding the phase differential vs frequencyresponse for each of the components and properly combining appropriatelysized components can be used to provide extremely good circular polarityperformance (e.g., low cross polarity levels) by maintaining 90 degreephase differential over significant frequency band width. In particular,the sizes can be chosen such that the phase differential (versusfrequency response) between orthogonal linear components introduced inthe coaxial and low frequency waveguides by the z distance d₅ betweenthe boards is oppositely sloped compared to the phase differential (vs.frequency response) introduced by the difference in trace lengths,specifically the difference in lengths of traces 2802 and 2804 comparedto the lengths of traces 2806 and 2808.

Some embodiments of this disclosure overcome the limited bandwidthperformance by greatly reducing the slope of the phase differential vsfrequency response, resulting in phase differential remaining close to90 degrees over a much broader frequency band. For example, this may beaccomplished by introducing a portion of the needed phase differentialby separating the traces for the LP components a distance d₅ (e.g.,approximately 0.04 inches) in the z direction, while making the lengthsof traces 2806 and 2808 that support LP component 2 longer (e.g., byapproximately 0.12 inches) than the lengths of traces 2802 and 2804 thatsupport orthogonal LP component 1, and by using a ridge polarizersection to introduce another portion of the phase differential. Thisdifference in trace length introduces a phase differential vs frequencyresponse that is oppositely sloped compared to the phase differentialintroduced by the ridge polarizer section. The result may be a totalphase differential vs frequency that has a relatively flat slope over avery wide band, remaining very close to 90 degrees, which will bydefinition, result in very good CP Xpol performance over a widefrequency band, as shown by graph 3200 in FIG. 32.

Another embodiment of the present disclosure overcomes the limitedbandwidth circular polarity performance and eliminates the need for theadditional CP polarity mechanism (ridges, septums etc.). In oneembodiment, this may be accomplished by both separating the traces forthe 2 orthogonal LP components a distance d₅ (e.g., approximately 0.25inches) in the z direction, while at the same time making the lengths oftraces 2806 and 2808 that support LP component 2 longer (e.g.,approximately 0.174 inches) than the lengths of traces 2802 and 2804that support orthogonal LP component 1. The phase differential vsfrequency introduced by the difference in trace lengths is oppositelysloped from that of the phase differential vs frequency introduced bythe distance d₅ as shown by graph 3300, in FIG. 33. The resulting totalphase differential vs frequency may have a relatively flat slop over avery wide band, remaining very close to 90 degrees. This may lead tovery good radiated CP Xpol performance over a wide frequency band whencombined with appropriated Horn radiating structures. For this examplethe CP Xpol performance is better than −34 dB over the entire desiredfrequency band providing excellent margin over the typical −20 to −25 dBCP X-pol that most satellite applications require.

It should be noted that reciprocity applies to passive systems so ifinstead a Low band transmitter replaced the LNB receiver then the twooutgoing linear polarities from the low band transmitter would beconverted to circular polarity and radiated out of the Horn.

In contrast to using conventional rectangular waveguide, the smallerchannel widths of the present disclosure allows the low wavelengthcombiner (e.g., the low wavelength combiner 110) to be considerablysmaller improving aesthetics and reducing cost. When formed into bendsor when interfaced with other structures (like the probes inside thecoaxial waveguide and circular waveguide), the inherent broad bandcharacteristics of these transmission lines make it easier to obtaingood return loss or match (in comparison to conventional waveguide),especially over wider frequency bands.

Conventional rectangular waveguides can be used in two possibleorientations. One orientation has the broad side of the waveguide in theplan parallel to the combiner which provides a relatively flatstructure. As a high volume low cost assembly this would be cast intoseveral layered casting with splits (mating surfaces) along the narrowwall of the wave guide where the castings would bolt together to formthe waveguides. This is problematic because gaps might occur at theseams significantly degrading performance by increasing loss, leakage,and possibly even creating unwanted resonances. If/when that happens itfundamentally changes the propagation characteristics of the waveguidetransmission line structure introducing unwanted discontinuities. Thisis particularly damaging for conventional rectangular waveguide giventhe orientation of the fields within it. The fields near the narrow wallare perpendicular to the broad wall and parallel to the narrow wall, sothe fields can easily enter gaps in the narrow wall (between the matingsurfaces of the cast pieces). When such gaps open up, the gapseffectively change the width of the waveguide structure increasing thefield concentrations near the narrow wall and inside the unwanted gap,where energy can leak or create unwanted resonances which can be verydamaging to performance. If other metal structures are present in closeproximity to the waveguide the energy could leak back into the waveguideout of phase creating resonances significantly increasing the insertionloss at some frequencies and drastically degrading performance.

It will be appreciated that variants of the above-disclosed and otherfeatures and functions, or alternatives thereof, may be combined intomany other different systems or applications. Various presentlyunforeseen or unanticipated alternatives, modifications, variations, orimprovements therein may be subsequently made by those skilled in theart which are also intended to be encompassed by the following claims.

What is claimed is:
 1. An apparatus, comprising: a low frequencyradiator; a high frequency radiator; a high frequency waveguide thatcarries high frequency bands to the high frequency radiator; a lowfrequency coaxial waveguide coupled to the high frequency waveguide in acoaxial structure, wherein the low frequency coaxial waveguide carrieslow frequency bands to the low frequency radiator; and a low frequencycombiner in communication with the low frequency coaxial waveguide,wherein the low frequency combiner comprises a circular low frequencywaveguide and air dielectric transmission lines formed by air channelsformed above and below a plurality of printed circuits in a metalhousing, wherein the air dielectric transmission lines comprise: a topmetal housing portion of the metal housing; a first printed circuit ofthe plurality of printed circuits; a middle housing portion of the metalhousing coupled to the top metal housing portion to form first airchannels above and below the first printed circuit within a volumeformed by the top metal housing portion and the middle housing portion;a second printed circuit of the plurality of printed circuits; and abottom metal housing portion of the metal housing coupled to the middlehousing portion to form second air channels above and below the secondprinted circuit within a volume formed by the middle housing portion andthe bottom metal housing portion.
 2. The apparatus of claim 1, furthercomprising a high frequency interface port in communication with thehigh frequency waveguide to receive the high frequency bands.
 3. Theapparatus of claim 1, wherein the low frequency combiner comprises a lowfrequency interface port in communication with the circular lowfrequency waveguide to receive the low frequency bands.
 4. The apparatusof claim 1, wherein the plurality of printed circuits are printed on aprinted circuit board (PCB) or a thin film.
 5. The apparatus of claim 4,wherein the PCB comprises micro-strip printed circuits or striplineprinted circuits.
 6. The apparatus of claim 1, wherein the first printedcircuit carries a first polarity of the low frequency bands and thesecond printed circuit carries a second polarity of the low frequencybands.
 7. The apparatus of claim 1, wherein the first printed circuitcomprises: a first metal trace; a second metal trace; a first probecoupled to a first end of the first metal trace; a second probe coupledto a second end of the first metal trace; a third probe coupled to afirst end of the second metal trace, wherein the third probe is locatedopposite the first probe; and a fourth probe coupled to a second end ofthe second metal trace, wherein the fourth probe is located opposite thesecond probe.
 8. The apparatus of claim 7, wherein a length of the firstmetal trace is equal to a length of the second metal trace.
 9. Theapparatus of claim 7, wherein the first metal trace and the second metaltrace are on a single side of the first printed circuit.
 10. Theapparatus of claim 7, wherein the first metal trace and the second metaltrace comprise duplicate traces on an opposing side of the first printedcircuit, wherein the first metal trace and the second metal trace areconnected to the duplicate traces via a plurality of platted throughvias.
 11. The apparatus of claim 10, wherein the plurality of plattedthrough vias align with an outer perimeter of the air channels thatcontain the first metal trace and the second metal trace.
 12. Theapparatus of claim 7, wherein the second printed circuit comprises: athird metal trace; a fourth metal trace; a fifth probe coupled to afirst end of the third metal trace; a sixth probe coupled to a secondend of the third metal trace a seventh probe coupled to a first end ofthe fourth metal trace, wherein the seventh probe is located oppositethe fifth probe; and an eighth probe coupled to a second end of thefourth metal trace, wherein the eighth probe is located opposite thesixth probe.
 13. The apparatus of claim 12, wherein a length of thethird metal trace is equal to a length of the fourth metal trace. 14.The apparatus of claim 12, wherein the third metal trace and the fourthmetal trace are on a single side of the second printed circuit.
 15. Theapparatus of claim 12, wherein the third metal trace and the fourthmetal trace comprise duplicate traces on an opposing side of the secondprinted circuit, wherein the third metal trace and the fourth metaltrace are connected to the duplicate traces via a plurality of plattedthrough vias.
 16. The apparatus of claim 15, wherein the plurality ofplatted through vias align with an outer perimeter of the air channelsthat contain the third metal trace and the fourth metal trace.
 17. Theapparatus of claim 12, wherein the first probe and the third probe arelocated approximately 90 degrees relative to the fifth probe and theseventh probe around a perimeter of a first opening and the second probeand the fourth probe are located approximately 90 degrees relative tothe sixth probe and the eighth probe around a perimeter of a secondopening, wherein the coaxial structure of the high frequency waveguideand the low frequency coaxial waveguide is located through the firstopening and the circular low frequency waveguide is located through thesecond opening.
 18. The apparatus of claim 1, wherein a length ofchannels within each one of the plurality of printed circuits, adistance along a z-axis between two or more of the plurality of printedcircuits, or a combination of the length and the distance, are set incombination with an additional circular polarizer mechanism comprisingat least one of: ridges, septums, oblong waveguides, or dielectric vanesto receive or to transmit circular polarity signals.
 19. The apparatusof claim 1, wherein a length of channels within each one of theplurality of printed circuits and a distance along a z-axis between twoor more of the plurality of printed circuits are set to receive or totransmit circular polarity signals.